FIG. 1 shows a prior art OFDM receiver 10. A baseband signal 12 enters a synchronization function 14, which serves to identify phase and frequency offsets in the incoming signal 12, where they are fed back to an NCO (not shown) or a phase rotator (not shown) which removes the offsets and frequency drifts from the synchronized signal. The phase and frequency corrected signal 15 is delivered to an FFT 16 which recovers the combinations of OFDM subcarriers which comprise the transmitted data. FFT outputs 17 are shown as signal 17a, comprising linear combinations of FFT output data having real and imaginary components. The FFT output 17 is provided to a channel estimation and equalization function 18, which produces output 19 compensated for channel phase and magnitude variations. Plot 19a shows the output 19 in a frequency vs real and imaginary amplitude view, and plot 19b shows the corresponding constellation diagram for 16-QAM, where each position in a 16 QAM constellation diagram represents 4 bits of data after decoding. The output 19 of the channel compensator 18 is fed to the soft constellation de-mapper 24, which performs the function of converting the constellation into corresponding data values, and this output 23 is fed to the de-interleaver and soft decoder 20, which performs data decoding resulting in output data 22.
FIG. 2 shows a preamble stream 25 for an OFDM packet. The packet 25 comprises a sequence of preamble tones P0 through P15 which form a first preamble 26 followed by a second identical preamble 28, which is followed by a third preamble 30, and finally the packet data 32. During the preamble times corresponding to preambles 26, 28, and 30 of packet 25, the synchronization function 14 and channel estimation function 18 of FIG. 1 make estimations of channel frequency offset, phase offset, and channel frequency transfer function, respectively.
FIG. 3 shows one implementation of a prior art packet detection and coarse frequency offset synchronizer such as 14 of FIG. 1. The synchronizer comprises two parts, a coarse frequency offset part 40, and a packet detection part 60. The frequency offset estimator 40 accepts as an input a stream of complex OFDM symbols 92 and a delayed version 42 of the same stream, where the delay is equal to the interval of a single preamble interval 26. The conjugator 52 has the function of inverting the imaginary part of the incoming stream such that a+jb becomes a−jb. The product of (a+jb)(a−jb) produces the signal power level a2+b2, since the same-position preamble symbols are identical other than the frequency offset generated phase shift component from the earlier symbol to the later symbol. Consequently, the multiplier 44 output contains an imaginary component corresponding to the amount of phase shift from a first preamble symbol to a second preamble symbol. The Phase Finder 46, which is implemented as a CORDIC generates an output 47 which represents the phase φ of the incoming multiplier 44 product. The frequency may be then be estimated from change of phase per sample Δφ/Δt. The output of CORDIC 46 is averaged 48 to generate a coarse frequency offset 50. This value is measured during the preamble interval and fed back to a numerically controlled oscillator (NCO, not shown) or phase rotator (not shown) to remove any frequency offset during the balance of the packet receive time prior to performing the FFT, where such frequency offset would result in an offset in the FFT 16 of FIG. 1 outputs.
The symbol timing may be extracted from the processing shown as packet detection system 60 of FIG. 3. The incoming stream of baseband OFDM symbols are delayed 62 by a time equal to a preamble interval, and the preamble stream 92 is multiplied 66 by a delayed preamble 63 and conjugated 64 to produce multiplier 66 output 67. This output 67 is averaged over an interval equal to the number of symbols in a preamble (shown as 16 symbols) to generate a value Cn 74, which represents the power level of the signal, as before. During the preamble interval, the multiplication of a current preamble symbol with the same symbol from a previous preamble results in the output 67 of the multiplier 66 representing the correlated signal power. The averager 70 sums the previous preamble values (shown for a 16 symbol preamble) to generate a power value Cn 74 whose value represents the noise plus interference component of the SINR value to be determined. The output 63 of the delay element 62 is multiplied by a conjugate 64 value 65 to produce a product 69, which is averaged over the same preamble interval by averager 72 to generate a signal plus noise power level 76. Since there is very little signal correlation from one symbol of a preamble to the next, the output Pn 76 provides an indication of the uncorrelated noise plus interference level, which includes unrelated noise and interference effects such as preamplifier gain in the RF signal processing chain and reflected signal energy, in contrast to the correlated value Cn 74 indicates the correlated power level of the incoming stream during the preamble interval. Cn 74 and Pn 76 are ordinarily used to establish the symbol timing referenced to the preamble, and one such method is to divide 78 the absolute value of Cn 84 by the noise plus signal level Pn 76 to generate a figure of merit μ 85, and to associate packet detection 90 with μ 85 crossing some predetermined threshold using a comparator 88.
FIG. 4 shows the signals for the prior art packet detection system of FIG. 3. The packet preamble is shown as 120, while signal power 67 is shown as 122 and noise and interference power signal 69 is shown as 124. Output Cn 74 is shown as signal 126, and output Pn 76 is shown as signal 128, which both rise during second preamble time t2, which corresponds to interval 28 of FIG. 2. The ratio of Cn/Pn is shown on waveform 127, and when waveform 127 crosses threshold 125, start of packet 121 is indicated, while end of preamble/start of data/symbol timing may be detected by falling correlated signal waveform 122 edge 123.
The use of existing signals Cn and Pn is known in the prior art for symbol timing and packet detection, and it also known in the prior art to change demodulation method and transmission speed based on error rate at the detector. It is desired to generate a SINR estimate using these signals for use in demodulation, particularly following the soft constellation demapping step, whereby the quantization method performed on the demapped data may be changed in accordance with the value of SINR as determined during the preamble synchronization step.
An estimate of the receiver signal quality can be used to improve the performance or reduce the complexity of base-band processing functions. An estimate of the noise variance is a sufficient measure of the signal quality, as the AGC (Automatic Gain Control) function of the RF receiver (not shown) ensures constant input power to a base-band system. Typically, symbol decisions are compared with the received symbol to obtain an error vector. The error vectors can be averaged to obtain an estimate of the noise variance as discussed in U.S. Pat. No. 5,379,324. The symbol decisions can be made at the input to the decoder, or at the decoder output. Using decisions from the output of the decoder provides a better estimate of the noise variance. Both these techniques have significant latency, and it is useful to have an estimate of signal strength established during the preamble interval so that it may be used during the data interval of the same packet. It is desired to have a signal strength estimation for use in an OFDM system which relies on parameters which can be established during the preamble interval.
A technique for synchronization based on a training sequence consisting of repeating patterns is described in “Robust Frequency and Timing Synchronization for OFDM”, IEEE Transactions on communications, December 1997. As noted in FIG. 3 and FIG. 4, due to the repeating preamble symbols, a correlation peak is observed at the end of the training sequence. This peak is used to detect a valid reception. The position of the peak also indicates the symbol boundary.
The correlation be represented as,
      C    ⁡          (      n      )        =            ∑                        n          -          L                <        k        ≤        n              ⁢                  X        ⁡                  (          k          )                    *                        X          ⁡                      (                          k              -              L                        )                          *            The signal energy is computed as,
      E    ⁡          (      n      )        =            1      2        ⁢                  ∑                              n            -                          2              ⁢              L                                <          k          ⁢                      <            _                    ⁢          n                    ⁢                          ⁢                                              X            ⁡                          (              k              )                                                2            
The normalized value used for symbol timing is given by
      Y    ⁡          (      n      )        =                                      C          ⁡                      (            n            )                                      2                      E        ⁡                  (          n          )                    2      
In an OFDM system, the soft metric values can be weighed by the corresponding channel estimates resulting in a significant improvement in receiver performance. A simple technique to generate soft metrics with channel weighting is discussed in “Simplified Soft-Output Demapper for Binary Interleaved COFDM with application to Hiperlan/2” by Tosato et al.
In a frequency selective fading environment, where OFDM is typically employed, the frequency domain channel estimates have large peaks and nulls and therefore a large dynamic range as shown in 148 FIG. 7. This requires a large soft metric bit-width to accurately represent the reliability information. A large soft-metric bit-width results in a significant increase in area as bit-widths in De-Interleaver and Soft decoder blocks increase. A non-uniform quantization technique could be used but it leads to increased complexity in the soft decoders.
A technique for soft metric quantization is discussed in U.S. Pat. No. 5,379,324. This technique uses statistical information from the output of the soft metric quantization to adjust the quantization threshold. This is however an iterative process with inherent latency. The proposed technique provides a simpler technique to determine the quantization threshold for OFDM systems without any latency.
U.S. Pat. No. 5,214,675 by Mueller et al. describes a system for compensating for multi-path reflection in a communications system by computing a variance of the signal and providing this signal to a filter which compensates for multipath delay.
U.S. Pat. No. 6,792,055 by Hart describes a system for use in QAM whereby the strength of the demodulated signal is fed back to a gain control. In another embodiment, the decoder makes hard and soft decisions according to a variable threshold which is set by the strength of the signal applied to the decoder.
U.S. Pat. No. 5,740,203 describes a prior art demapper for QAM and PSK modulation methods which performs the function of block 24 of FIG. 1 or block 140 of FIG. 6.
U.S. Pat. No. 5,379,324 by Mueller et al describes a system for computing gain and noise variance of a channel for use in correcting the channel.